Capacitance measurement

ABSTRACT

A differential amplifier has an output and differential first and second inputs. A switch disposed between a sensor electrode and the second input is opened to initiate a reset phase where the sensor electrode and the differential amplifier are decoupled. A feedback capacitance disposed between the second input and the output is reset to a first level of charge. The switch is closed to initiate a measurement phase where the second input and sensor electrode are coupled. In the measurement phase: charge is balanced between the sensor electrode and the feedback capacitance such that a sensor electrode voltage equals a voltage of the first input equals a voltage of the second input, and the sensor electrode is charged; and the differential amplifier is utilized to integrate charge on the sensor electrode, such that an absolute capacitance corresponding to a coupling between the sensor electrode and an input object is measured.

CROSS-REFERENCE TO RELATED U.S. PROVISIONAL PATENT APPLICATIONS

This application claims priority to and benefit of U.S. ProvisionalPatent Application No. 61/673,241 filed on Jul. 18, 2012 entitled“SYSTEM AND METHOD FOR SENSING ABSOLUTE CAPACITANCE” by FarzanehShahrokhi et al., and assigned to the assignee of the presentapplication.

This application claims priority to and benefit of U.S. ProvisionalPatent Application No. 61/693,541 filed on Aug. 27, 2012 entitled“SYSTEM AND METHOD FOR SENSING ABSOLUTE CAPACITANCE” by FarzanehShahrokhi et al., and assigned to the assignee of the presentapplication.

BACKGROUND

Input devices including proximity sensor devices (also commonly calledtouchpads or touch sensor devices) are widely used in a variety ofelectronic systems. A proximity sensor device typically includes asensing region, often demarked by a surface, in which the proximitysensor device determines the presence, location and/or motion of one ormore input objects. Proximity sensor devices may be used to provideinterfaces for the electronic system. For example, proximity sensordevices are often used as input devices for larger computing systems(such as opaque touchpads integrated in, or peripheral to, notebook ordesktop computers). Proximity sensor devices are also often used insmaller electronic devices/systems (such as touch screens integrated incellular phones and tablet computers). Such touch screen input devicesare typically superimposed upon or otherwise collocated with a displayof the electronic device/system.

SUMMARY

In a method of capacitance measurement with a differential amplifierhaving an output and differential first and second inputs, a switchdisposed between a sensor electrode and the second input of thedifferential amplifier is opened to initiate a reset phase where thesensor electrode and the differential amplifier are decoupled. Afeedback capacitance disposed between the second input and the output isreset to a first level of charge. The switch is closed to initiate ameasurement phase where the second input and the sensor electrode arecoupled. In the measurement phase: charge is balanced between the sensorelectrode and the feedback capacitance such that a sensor electrodevoltage equals a voltage of the first input equals a voltage of thesecond input, and the sensor electrode is charged to a valueproportional to its capacitance and the voltage of the second input; andthe differential amplifier is utilized to integrate charge on the sensorelectrode, such that an absolute capacitance corresponding to a couplingbetween the sensor electrode and an input object is measured.

BRIEF DESCRIPTION OF DRAWINGS

The drawings referred to in this Brief Description of Drawings shouldnot be understood as being drawn to scale unless specifically noted. Theaccompanying drawings, which are incorporated in and form a part of theDescription of Embodiments, illustrate various embodiments and, togetherwith the Description of Embodiments, serve to explain principlesdiscussed below, where like designations denote like elements.

FIG. 1 is a block diagram of an example input device, in accordance withembodiments.

FIG. 2 shows a portion of an example sensor electrode pattern which maybe utilized in a sensor to generate all or part of the sensing region ofan input device, such as a touch screen, according to some embodiments.

FIG. 3 shows a comparison of diagrams of transcapacitive andconventional absolute capacitance sensing signals and modes.

FIGS. 4A and 4B illustrate operation of a capacitance measurementcircuit through a first half cycle of absolute capacitive sensing,according to an embodiment.

FIG. 5 shows a comparison of diagrams of transcapacitive and new (asdescribed herein) absolute capacitance sensing signals and mode, inaccordance with various embodiments.

FIGS. 6A-6D illustrate operation of an capacitive measurement circuitthrough a full cycle of absolute capacitive sensing, according to anembodiment.

FIG. 7 illustrates a capacitance measurement circuit, according to anembodiment.

FIG. 8 illustrates a capacitance measurement circuit, according to someembodiments.

FIGS. 9A-9C illustrate a method of capacitance measurement with adifferential amplifier having an output and differential first andsecond input, according to various embodiments.

DESCRIPTION OF EMBODIMENTS

The following Description of Embodiments is merely provided by way ofexample and not of limitation. Furthermore, there is no intention to bebound by any expressed or implied theory presented in the precedingBackground, Summary, or Brief Description of Drawings or the followingDescription of Embodiments.

Overview of Discussion

Herein, various embodiments are described that provide input devices,processing systems, and methods that facilitate improved usability. Invarious embodiments described herein, the input device may be acapacitive input device.

Discussion begins with a comparison of transcapacitive sensing signalsand modes with conventional absolute capacitance sensing signals andmodes. In accordance with new embodiments described herein, an examplecapacitance measurement circuit is then presented and its operationdescribed. A comparison of transcapacitive sensing signals and modeswith signals and modes of new embodiments of absolute capacitive sensingis then presented and described. Several variations to the constructionand operation of a new absolute capacitance measurement circuit are thenpresented and described. A hybrid transcapacitive/absolute capacitancemeasurement circuit is then presented and described. Operation ofvarious new capacitive measurement circuits is then further described inconjunction with description of a method of capacitance measurement witha differential amplifier having an output and differential first andsecond input.

Example Input Device

Turning now to the figures, FIG. 1 is a block diagram of an exemplaryinput device 100, in accordance with various embodiments. Input device100 may be configured to provide input to an electronic system/device(not depicted). As used in this document, the term “electronic system”(or “electronic device”) broadly refers to any system capable ofelectronically processing information. Some non-limiting examples ofelectronic systems include personal computers of all sizes and shapes,such as desktop computers, laptop computers, netbook computers, tablets,web browsers, e-book readers, and personal digital assistants (PDAs).Additional example electronic systems include composite input devices,such as physical keyboards that include input device 100 and separatejoysticks or key switches. Further example electronic systems includeperipherals such as data input devices (including remote controls andmice), and data output devices (including display screens and printers).Other examples include remote terminals, kiosks, and video game machines(e.g., video game consoles, portable gaming devices, and the like).Other examples include communication devices (including cellular phones,such as smart phones), and media devices (including recorders, editors,and players such as televisions, set-top boxes, music players, digitalphoto frames, and digital cameras). Additionally, the electronic systemscould be a host or a slave to the input device.

Input device 100 can be implemented as a physical part of an electronicsystem, or can be physically separate from an electronic system. Asappropriate, input device 100 may communicate with parts of theelectronic system using any one or more of the following: buses,networks, and other wired or wireless interconnections. Examplesinclude, but are not limited to: Inter-Integrated Circuit (I2C), SerialPeripheral Interface (SPI), Personal System 2 (PS/2), Universal SerialBus (USB), Bluetooth®, Radio Frequency (RF), and Infrared DataAssociation (IrDA).

In FIG. 1, input device 100 is shown as a proximity sensor device (alsooften referred to as a “touchpad” or a “touch sensor device”) configuredto sense input provided by one or more input objects 140 in a sensingregion 120. Example input objects include fingers and styli, as shown inFIG. 1.

Sensing region 120 encompasses any space above, around, in and/or nearinput device 100, in which input device 100 is able to detect user input(e.g., user input provided by one or more input objects 140). The sizes,shapes, and locations of particular sensing regions may vary widely fromembodiment to embodiment. In some embodiments, sensing region 120extends from a surface of input device 100 in one or more directionsinto space until signal-to-noise ratios prevent sufficiently accurateobject detection. The distance to which this sensing region 120 extendsin a particular direction, in various embodiments, may be on the orderof less than a millimeter, millimeters, centimeters, or more, and mayvary significantly with the type of sensing technology used and theaccuracy desired. Thus, some embodiments sense input that comprises nocontact with any surfaces of input device 100, contact with an inputsurface (e.g., a touch surface) of input device 100, contact with aninput surface of input device 100 coupled with some amount of appliedforce or pressure, and/or a combination thereof. In various embodiments,input surfaces may be provided by surfaces of casings within which thesensor electrodes reside, by face sheets applied over the sensorelectrodes or any casings, etc. In some embodiments, sensing region 120has a rectangular shape when projected onto an input surface of inputdevice 100.

Input device 100 may utilize any combination of sensor components andsensing technologies to detect user input in sensing region 120. Inputdevice 100 comprises one or more sensing elements for detecting userinput. As a non-limiting example, input device 100 may use capacitivetechniques.

Some implementations are configured to provide images that span one,two, three, or higher dimensional spaces. Some implementations areconfigured to provide projections of input along particular axes orplanes.

In some capacitive implementations of input device 100, voltage orcurrent is applied to create an electric field. Nearby input objectscause changes in the electric field, and produce detectable changes incapacitive coupling that may be detected as changes in voltage, current,or the like.

Some capacitive implementations utilize arrays or other regular orirregular patterns of capacitive sensing elements to create electricfields. In some capacitive implementations, separate sensing elementsmay be ohmically shorted together to form larger sensor electrodes. Somecapacitive implementations utilize resistive sheets, which may beuniformly resistive.

Some capacitive implementations utilize “self capacitance” (or “absolutecapacitance”) sensing methods based on changes in the capacitivecoupling between sensor electrodes and an input object. In variousembodiments, an input object near the sensor electrodes alters theelectric field near the sensor electrodes, thus changing the measuredcapacitive coupling. In one implementation, an absolute capacitancesensing method operates by modulating sensor electrodes with respect toa reference voltage (e.g., system ground), and by detecting thecapacitive coupling between the sensor electrodes and input objects.

Some capacitive implementations utilize “mutual capacitance” (or“transcapacitance”) sensing methods based on changes in the capacitivecoupling between sensor electrodes. In various embodiments, an inputobject near the sensor electrodes alters the electric field between thesensor electrodes, thus changing the measured capacitive coupling. Inone implementation, a transcapacitive sensing method operates bydetecting the capacitive coupling between one or more transmitter sensorelectrodes (also “transmitter electrodes” or “transmitters”) and one ormore receiver sensor electrodes (also “receiver electrodes” or“receivers”). Collectively transmitters and receivers may be referred toas sensor electrodes or sensor elements. Transmitter sensor electrodesmay be modulated relative to a reference voltage (e.g., system ground)to transmit transmitter signals. Receiver sensor electrodes may be heldsubstantially constant relative to the reference voltage to facilitatereceipt of resulting signals. A resulting signal may comprise effect(s)corresponding to one or more transmitter signals, and/or to one or moresources of environmental interference (e.g., other electromagneticsignals). Sensor electrodes may be dedicated transmitters or receivers,or may be configured to both transmit and receive. In some embodiments,one or more receiver electrodes may be operated to receive a resultingsignal when no transmitter electrodes are transmitting (e.g., thetransmitters are disabled). In this manner, the resulting signalrepresents noise detected in the operating environment of sensing region120.

In FIG. 1, a processing system 110 is shown as part of input device 100.Processing system 110 is configured to operate the hardware of inputdevice 100 to detect input in sensing region 120. Processing system 110comprises parts of or all of one or more integrated circuits (ICs)and/or other circuitry components. (For example, a processing system fora mutual capacitance sensor device may comprise transmitter circuitryconfigured to transmit signals with transmitter sensor electrodes,and/or receiver circuitry configured to receive signals with receiversensor electrodes). In some embodiments, processing system 110 alsocomprises electronically-readable instructions, such as firmware code,software code, and/or the like. In some embodiments, componentscomposing processing system 110 are located together, such as nearsensing element(s) of input device 100. In other embodiments, componentsof processing system 110 are physically separate with one or morecomponents close to sensing element(s) of input device 100, and one ormore components elsewhere. For example, input device 100 may be aperipheral coupled to a desktop computer, and processing system 110 maycomprise software configured to run on a central processing unit of thedesktop computer and one or more ICs (perhaps with associated firmware)separate from the central processing unit. As another example, inputdevice 100 may be physically integrated in a phone, and processingsystem 110 may comprise circuits and firmware that are part of a mainprocessor of the phone. In some embodiments, processing system 110 isdedicated to implementing input device 100. In other embodiments,processing system 110 also performs other functions, such as operatingdisplay screens, driving haptic actuators, etc.

Processing system 110 may be implemented as a set of modules that handledifferent functions of processing system 110. Each module may comprisecircuitry that is a part of processing system 110, firmware, software,or a combination thereof. In various embodiments, different combinationsof modules may be used. Example modules include hardware operationmodules for operating hardware such as sensor electrodes and displayscreens, data processing modules for processing data such as sensorsignals and positional information, and reporting modules for reportinginformation. Further example modules include sensor operation modulesconfigured to operate sensing element(s) to detect input, identificationmodules configured to identify gestures such as mode changing gestures,and mode changing modules for changing operation modes.

In some embodiments, processing system 110 responds to user input (orlack of user input) in sensing region 120 directly by causing one ormore actions. Example actions include changing operation modes, as wellas GUI actions such as cursor movement, selection, menu navigation, andother functions. In some embodiments, processing system 110 providesinformation about the input (or lack of input) to some part of theelectronic system (e.g., to a central processing system of theelectronic system that is separate from processing system 110, if such aseparate central processing system exists). In some embodiments, somepart of the electronic system processes information received fromprocessing system 110 to act on user input, such as to facilitate a fullrange of actions, including mode changing actions and GUI actions.

For example, in some embodiments, processing system 110 operates thesensing element(s) of input device 100 to produce electrical signalsindicative of input (or lack of input) in sensing region 120. Processingsystem 110 may perform any appropriate amount of processing on theelectrical signals in producing the information provided to theelectronic system. For example, processing system 110 may digitizeanalog electrical signals obtained from the sensor electrodes. Asanother example, processing system 110 may perform filtering or othersignal conditioning. As yet another example, processing system 110 maysubtract or otherwise account for a baseline, such that the informationreflects a difference between the electrical signals and the baseline.As yet further examples, processing system 110 may determine positionalinformation, recognize inputs as commands, recognize handwriting, andthe like.

“Positional information” as used herein broadly encompasses absoluteposition, relative position, velocity, acceleration, and other types ofspatial information. Exemplary “zero-dimensional” positional informationincludes near/far or contact/no contact information. Exemplary“one-dimensional” positional information includes positions along anaxis. Exemplary “two-dimensional” positional information includesmotions in a plane. Exemplary “three-dimensional” positional informationincludes instantaneous or average velocities in space. Further examplesinclude other representations of spatial information. Historical dataregarding one or more types of positional information may also bedetermined and/or stored, including, for example, historical data thattracks position, motion, or instantaneous velocity over time.

In some embodiments, input device 100 is implemented with additionalinput components that are operated by processing system 110 or by someother processing system. These additional input components may provideredundant functionality for input in sensing region 120, or some otherfunctionality. FIG. 1 shows buttons 130 near sensing region 120 that canbe used to facilitate selection of items using input device 100. Othertypes of additional input components include sliders, balls, wheels,switches, and the like. Conversely, in some embodiments, input device100 may be implemented with no other input components.

In some embodiments, input device 100 may be a touch screen, and sensingregion 120 overlaps at least part of an active area of a display screen.For example, input device 100 may comprise substantially transparentsensor electrodes overlaying the display screen and provide a touchscreen interface for an associated electronic system. The display screenmay be any type of dynamic display capable of displaying a visualinterface to a user, and may include any type of light emitting diode(LED), organic LED (OLED), cathode ray tube (CRT), liquid crystaldisplay (LCD), plasma, electroluminescence (EL), or other displaytechnology. Input device 100 and the display screen may share physicalelements. For example, some embodiments may utilize some of the sameelectrical components for displaying and sensing. As another example,the display screen may be operated in part or in total by processingsystem 110.

It should be understood that while many embodiments are described in thecontext of a fully functioning apparatus, the mechanisms are capable ofbeing distributed as a program product (e.g., software) in a variety offorms. For example, the mechanisms that are described may be implementedand distributed as a software program on information bearing media thatare readable by electronic processors (e.g., non-transitorycomputer-readable and/or recordable/writable information bearing mediareadable by processing system 110). Additionally, the embodiments applyequally regardless of the particular type of medium used to carry outthe distribution. Examples of non-transitory, electronically readablemedia include various discs, memory sticks, memory cards, memorymodules, and the like. Electronically readable media may be based onflash, optical, magnetic, holographic, or any other tangible storagetechnology.

Example Sensor Electrode Pattern

FIG. 2 shows a portion of an example sensor electrode pattern 200 whichmay be utilized in a sensor to generate all or part of the sensingregion of a input device 100, according to various embodiments. Inputdevice 100 is configured as a capacitive input device when utilized witha capacitive sensor electrode pattern. For purposes of clarity ofillustration and description, a non-limiting simple rectangular sensorelectrode pattern 200 is illustrated. It is appreciated that numerousother sensor electrode patterns may be employed including patterns witha single set of sensor electrodes, patterns with two sets of sensorelectrodes disposed in a single layer (without overlapping), andpatterns that provide individual button electrodes. The illustratedsensor electrode pattern is made up of a plurality of receiverelectrodes 270 (270-0, 270-1, 270-2 . . . 270-n) and a plurality oftransmitter electrodes 260 (260-0, 260-1, 260-2 . . . 260-n) whichoverlay one another, in this example. In the illustrated example, touchsensing pixels are centered at locations where transmitter and receiverelectrodes cross. Capacitive pixel 290 illustrates one of the capacitivepixels generated by sensor electrode pattern 200 during transcapacitivesensing. It is appreciated that in a crossing sensor electrode pattern,such as the illustrated example, some form of insulating material orsubstrate is typically disposed between transmitter electrodes 260 andreceiver electrodes 270. However, in some embodiments, transmitterelectrodes 260 and receiver electrodes 270 may be disposed on the samelayer as one another through use of routing techniques and/or jumpers.In various embodiments, touch sensing includes sensing input objectsanywhere in sensing region 120 and may comprise: no contact with anysurfaces of the input device 100, contact with an input surface (e.g., atouch surface) of the input device 100, contact with an input surface ofthe input device 100 coupled with some amount of applied force orpressure, and/or a combination thereof.

When accomplishing transcapacitive measurements, capacitive pixels, suchas capacitive pixel 290, are areas of localized capacitive couplingbetween transmitter electrodes 260 and receiver electrodes 270. Thecapacitive coupling between transmitter electrodes 260 and receiverelectrodes 270 changes with the proximity and motion of input objects inthe sensing region associated with transmitter electrodes 260 andreceiver electrodes 270.

In some embodiments, sensor electrode pattern 200 is “scanned” todetermine these capacitive couplings. That is, the transmitterelectrodes 260 are driven to transmit transmitter signals. Transmittersmay be operated such that one transmitter electrode transmits at onetime, or multiple transmitter electrodes transmit at the same time.Where multiple transmitter electrodes transmit simultaneously, thesemultiple transmitter electrodes may transmit the same transmitter signaland produce an effectively larger transmitter electrode, or thesemultiple transmitter electrodes may transmit different transmittersignals. For example, multiple transmitter electrodes may transmitdifferent transmitter signals according to one or more coding schemesthat enable their combined effects on the resulting signals of receiverelectrodes 270 to be independently determined.

The receiver electrodes 270 may be operated singly or multiply toacquire resulting signals. The resulting signals may be used todetermine measurements of the capacitive couplings at the capacitivepixels.

A set of measurements from the capacitive pixels form a “capacitiveimage” (also “capacitive frame”) representative of the capacitivecouplings at the pixels. Multiple capacitive images may be acquired overmultiple time periods, and differences between them used to deriveinformation about input in the sensing region. For example, successivecapacitive images acquired over successive periods of time can be usedto track the motion(s) of one or more input objects entering, exiting,and within the sensing region.

In some embodiments, one or more sensor electrodes 260 or 270 may beoperated to perform absolute capacitive sensing at a particular instanceof time. For example, receiver electrode 270-0 may be charged and thenthe capacitance of receiver electrode 270-0 may be measured. In such anembodiment, an input object 140 interacting with receiver electrode270-0 alters the electric field near receiver electrode 270-0, thuschanging the measured capacitive coupling. In this same manner, aplurality of sensor electrodes 270 may be used to measure absolutecapacitance and/or a plurality of sensor electrodes 260 may be used tomeasure absolute capacitance. It should be appreciated that whenperforming absolute capacitance measurements the labels of “receiverelectrode” and “transmitter electrode” lose the significance that theyhave in transcapacitive measurement techniques, and instead a sensorelectrode 260 or 270 may simply be referred to as a “sensor electrode.”

Comparison of Transcapacitive Sensing and Conventional AbsoluteCapacitive Sensing

FIG. 3 shows a comparison of diagrams of transcapacitive andconventional absolute capacitance sensing signals and modes (310 and 320respectively). FIG. 3, portion 310 illustrates transmitter andintegrated resulting signals (V_(TransInput) and V_(TransOut)respectively) for a transcapacitive sensing mode, where 311 is a firsthalf of a transcapacitive sensing cycle and 312 is a second half of atranscapacitive sensing cycle. Each half transcapacitive sensing cycle(311, 312) comprises an integration time period, T_(TransIntegrate), anda reset time period, T_(TransReset). In FIG. 3, portion 320 illustratestransmitter and integrated resulting signals (V_(ConvAbsInput) andV_(ConvAbsOut) respectively) for an absolute capacitive sensing modewherein 321 is the first half of a conventional absolute sensing cycleand 322 is the second half of the conventional absolute capacitivesensing cycle. Each half (321, 322) comprises one pre-charge timeperiod, T_(AbsPrecharge), and one integration time period,T_(AbsIntegrate). In the illustrated embodiments, the transmitter andresulting signals differ for the two sensing modes. For example, theillustrated conventional absolute capacitive sensing mode comprises apre-charge phase, T_(AbsPrecharge), in which a sensor electrode ischarged up to “voltage high” by the transmitter signal followed by anintegration phase, T_(AbsIntegrate), in which the sensor electrode isdischarged and the resulting charge flow from the resulting signal isintegrated and measured. For such an embodiment, the maximum amount ofcharge that may be measured by a capacitive measuring circuit coupled tothe sensor electrode is C_(B)(V_(dd)/2) where C_(B) is the absolutecapacitance (background capacitance+any input object capacitance) beingmeasured and V_(dd) is the receiver supply voltage (reference voltage oran operating voltage). For example, in this conventional absolutecapacitive sensing operation the pre-charge phase, T_(AbsPrecharge), andthe integration phase, T_(AbsIntegrate), durations may be based on thesettling time of the sensor electrode, T_(abs). This settling timeprecludes shortening these times without impact on the ability to sense.In many embodiments, as can be seen in FIG. 3, the transcapacitivesensing reset phase, T_(TransReset), is much shorter than theconventional absolute sensing pre-charge (T_(AbsPrecharge)) or absolutesensing integrate phase (T_(AbsIntegrate)) which is less than thetranscapacitive sensing integrate time,T_(TransReset)<<T_(aAbsPrecharge)<T_(TransInegrate). Therefore, theduration of a half sensing cycle for the conventional absolutecapacitive sensing method, T_(AbsPrecharge)+T_(AbsIntegrate), istypically greater than the duration of a half sensing cycle fortranscapacitive sensing method, T_(TransReset)+T_(TransIntegrate).Further, since the half sensing cycle for the conventional absolutecapacitive sensing method is greater than the half sensing cycle fortranscapacitive sensing method, the transmitter signal frequency forabsolute capacitive sensing is lower than the transmitter signalfrequency for transcapacitive sensing.

Example Capacitive Charge Measuring Circuits

As will be further described herein, in various embodiments, sensingfrequency of an absolute capacitive sensing device may be improved(shortened), as compared to conventional techniques, by alteringparameters of circuits and techniques used for performing absolutecapacitive sensing. For an absolute capacitive sensing device describedherein, increasing the amplitude and/or the frequency of transmittersignal may improve performance of the sensing device in comparison toconventional techniques of absolute capacitive sensing. For example, thesignal-to-noise ratio may be increased, interference susceptibility maybe improved and proximity sensing (distance and accuracy) may beimproved by one or more of increasing the amplitude and/or frequency ofthe transmitter signal. In various embodiments, increasing the amplitudeof the absolute sensing transmitter signal increases the proximitysensing distance and accuracy. Further, a capacitive sensing transmittersignal, such as V_(AbsInput) of FIG. 5, having a higher frequency mayincrease the avoidance of lower frequency interference components. Inyet other embodiments, an input device that is configured to operatewith an absolute capacitance measurement transmitter signal, such asV_(AbsInput) of FIG. 5, having increased amplitude and/or frequency (ascompared with conventional techniques for absolute capacitive sensing),may be configured to operate with a transmitter signal similar to thatof a transcapacitive sensing device. Such embodiments allow for theinterference susceptibility to be substantially the same for both modesof capacitive sensing, thus allowing interference avoidance for the twodifferent sensing modes can be coordinated. Further, an input deviceconfigured to operate in both a transcapacitive sensing mode and anabsolute capacitive sensing mode may be referred to as a hybridcapacitive sensor device. In such embodiments, the absolute capacitivesensing transmitter signal frequency may be at least equal to, if notfaster than, that of a transcapacitive transmitter signal. In oneembodiment, as is depicted in FIG. 5, the half sensing cycle for anabsolute capacitive sensing method may be at least equal to, if notfaster than, that of a transcapacitive sensing method. This will bedescribed in more detail below.

A hybrid capacitive sensing device may be configured to operate in botha transcapacitive sensing mode and an absolute capacitive sensing mode.In one embodiment, the hybrid capacitive sensing device is configured toswitch between a transcapacitive sensing mode and an absolute capacitivesensing mode based on, but not limited to, an operating state of theinput device, an input object event and a time delay. By way of exampleand not of limitation, in one embodiment, an absolute capacitive sensingmode may be used to detect the presence of an input object above, butnot touching an input surface of an input device, and in response todetection of such an input object the input device may switch from anabsolute capacitive sensing mode to a transcapacitive sensing mode.

In various embodiments, the amplitude and frequency of the absolutecapacitive sensing transmitter signal, V_(AbsInput), may be increased incomparison to the sensing frequency that is possible using conventionalabsolute capacitive sensing. In such embodiments, the reference voltage(operating voltage) of the charge integrator of the capacitancemeasuring circuit coupled to the sensor electrode may be modulated. Thereference voltage may be modulated symmetrically above and below areference value. The frequency and/or amplitude of the modulation may beadjusted during operation, such as to avoid interference, preventsaturation, or adjust the dynamic range of an amplifier. For example,the reference voltage may be modulated with signal that is similar tothe transcapacitive sensing signal (e.g., V_(TransInput)) in waveformand/or frequency (for example a square wave with similar frequency). Insuch an embodiment, the reference voltage, V_(ref), may be an attenuatedversion of the transcapacitive sensing signal. In other embodiments, thereference voltage may be modulated based on a receiver module supplyvoltage. In such an embodiment, the reference voltage, V_(ref), may bean attenuated version of the receiver module supply voltage, such as ½V_(dd). Further, in various embodiments, the reference voltage may bevariable. In other embodiments, the reference voltage may be selectedand configured to increase the dynamic range of the differentialamplifier being used as a charge integrator. In another embodiment, thereference voltage and the feedback capacitance may be selected toincrease the dynamic range of the differential amplifier being used as acharge integrator. Further, in other embodiments, the differentialamplifier being used as a charge integrator may be decoupled from thesensor electrode or electrodes which are being used for absolutecapacitive sensing during a reset phase of the charge integrator. In oneembodiment, decoupling the charge integrator from the sensor electrodeduring the reset phase provides a shorter reset phase for the chargeintegrator than would be experienced if it were to remain coupled to thesensor electrode during the reset phase. In some embodiments, the samecharge integrator may be configured to receive resulting signals whilean input device operates in a transcapacitive sensing mode and anabsolute capacitive sensing mode. Further, in some embodiments, thetranscapacitive sensing mode and the absolute sensing mode may beconfigured to use similar transmitter signals (similar in at least oneof frequency and amplitude). In some such embodiments, transmitterfrequency and the sensing cycles for the transcapacitive sensing modeand the absolute sensing mode are configured to be substantially thesame. For example, the absolute capacitive sensing “pre-charge/reset”and “integrate” durations, when combined are substantially equal to thereset and integrate times for transcapacitive sensing. The followingexample embodiments, describes various ways to provide an absolutecapacitive sensing having a reduced sensing cycle and an increasedtransmitter signal frequency.

FIGS. 4A and 4B illustrate operation of a capacitance measurementcircuit 400 through a first half cycle of absolute capacitive sensing,according to an embodiment. Capacitance measurement circuit 400 may beincluded as part of an input device 100 and/or a processing system 110.For example, processing system 110 may supply input voltages for circuit400 as well as control signals which operate switches in circuit 400and/or select capacitors from a bank of selectable capacitors. In FIGS.4A and 4B, capacitance measurement circuit 400 comprises a differentialamplifier 401 with inverting and non-inverting inputs and an output. Afirst switch SW1 is coupled between the non-inverting input ofdifferential amplifier 401 and a sensor electrode, such as sensorelectrode 270-0, to which circuit 400 is coupled. Differential amplifier401 is configured as a charge integrator and includes a feedbackcapacitance disposed between its output and its inverting input. Thefeedback capacitance is represented by capacitor, C_(FB), that iscoupled on one side to the output of differential amplifier 401 and onits other side to a location between the inverting input of differentialamplifier 401 and switch SW1. It should be appreciated that capacitorC_(FB) can be composed of one or more selectable capacitances that areselected from bank(s) of selectable capacitors. A second switch, SW2 isdisposed in parallel with feedback capacitor C_(FB). Switch SW2 operatesas a reset mechanism to discharge and reset capacitor C_(FB). In FIGS.4A and 4B, capacitance CB represents a background capacitance (which mayinclude capacitance contributed by an input object) between sensorelectrode 270-0 and ground.

Capacitance measurement circuit 400 performs an absolute sensing methodhaving a reduced half sensing cycle (as compared to conventionalabsolute capacitive sensing cycles), where differential amplifier 401 isset up as a charge integrator and the reference voltage applied to thenon-inverting input of differential amplifier 401 is modulated bysubstantially equal amounts above and below a reference voltage V_(ref).In some embodiments, the reference voltage of the integrating amplifieris modulated with a similar signal to the transcapacitive transmittersignal. In some embodiments, V_(ref) is approximately one half of asupply voltage, VDD. In such an embodiment, the absolute capacitivesensing frequency may be increased by using the propensity ofdifferential amplifier 401 to balance voltages on its inputs to drive atransmitter signal onto sensor electrode 270-0 which is equal to themodulated voltage applied on the non-inverting input. This method ofdriving is different than conventional absolute capacitive sensing whichuses a transmitter that is separate from a charge integrator.

FIG. 4A illustrates the reset phase of the first half sensing cycle.During this reset phase, the feedback capacitor, C_(FB), is dischargedand V_(out) follows the V_(inp) reference voltage, V_(AbsInput), that ismodulated onto the non-inverting input of differential amplifier 401. Inthe reset phase: switch SW2 is closed and switch SW1 is opened; and thepolarity of the modulation of V_(AbsInput) switches. The shift inreference voltage, V_(AbsInput), can occur at the very beginning of thereset phase or at some time after the reset phase has started. In someembodiments, when switch SW1 is open, sensor electrode 270-1 is left toelectrically float and thus substantially maintains whatever chargeremained upon it. Closing switch SW2 causes capacitor C_(FB) todischarge. The output and inverting input then take some time to settleto the new value of V_(AbsInput). The length of reset phase ispredicated upon how long the settling and discharge take to occur. Thedischarge of C_(FB) is fairly quick as there is no resistance in thepath to slow it, and the reset time of differential amplifier 401 ishastened by opening SW1 to disconnect it from sensor electrode 270-0 sothat it can more quickly settle without the capacitance of sensorelectrode 270-0 coupled to its inverting input.

FIG. 4B illustrates the integration phase of the first half sensingcycle. The integration phases are time periods where measurement takesplace and may also be referred to as measurement phases. In thisintegration phase, the non-inverting node of differential amplifier 401continues to be driven to the same value of V_(AbsInput) as during thereset phase shown in FIG. 4A. At the beginning of the integration phaseC_(FB) is discharged and both inputs and the output of differentialamplifier 401 have substantially settled at the voltage V_(AbsInput)being applied to the non-inverting input. To initiate the integrationphase, switch SW1 is closed and switch SW2 is opened. This causes thesensor electrode (e.g., 270-0) to be connected to the inverting input ofintegrating differential amplifier 401, which in-turn causes the voltageon background capacitance, C_(B), to be driven from the previous voltagelevel that it maintained after being decoupled from circuit 400 to thenew voltage that is present on the inverting input of differentialamplifier 401. For example in one integration cycle, C_(B) would bedriven from an old voltage of V_(ref)−ΔV_(ref) to a new voltage ofV_(ref)+ΔV_(ref), while in the following integration cycle on the nexthalf sensing cycle C_(B) would be driven from an old voltage ofV_(ref)+ΔV_(ref) to a new voltage of V_(ref)+ΔV_(ref). Thus, a charge of2ΔV_(ref) CB flows into C_(B) from C_(FB) and causes V_(out) to changefrom V_(ref)±ΔV_(ref) by 2ΔV_(ref) C_(B)/C_(FB).

FIG. 5 shows a comparison of diagrams of transcapacitive and new (asdescribed herein) absolute capacitance sensing signals and modes (310and 520 respectively), in accordance with various embodiments. Portion310 is the same as portion 310 of FIG. 3. Portion 520 illustrates a fullabsolute capacitive sensing cycle, according to embodiments describedherein in FIGS. 4A, 4B, 6A-6D, and FIG. 7. In FIG. 5, portion 520illustrates transmitter and integrated resulting signals (V_(AbsInput)and V_(out) respectively) for an absolute capacitive sensing modewherein 521 is the first half of a sensing cycle and 522 is the secondhalf of a capacitive sensing cycle. Each half sensing cycle 521 or 522includes both a pre-charge/reset phase, T_(AbsPrechargeReset), and anintegration phase, T_(AbsInegrate). The integration phases are timeperiods where measurement takes place and may also be referred to asmeasurement phases. The pre-charge/reset phase may include only reset(as illustrated in FIGS. 4A and 4B), some combination of reset andpre-charge (as illustrated in FIGS. 6A-6D), or only pre-charge asillustrated in FIG. 7. As can be seen, absolute capacitive measurementis accomplished with a reduced half sensing cycle period (as compared toconventional techniques of absolute capacitive sensing illustrated inFIG. 3). Although the pre-charge/reset phase is shown as beingsubstantially the same length of time as the integration phase wheremeasurement takes place, it should be appreciated that in someembodiments described herein the pre-charge/reset phase is shorter thanthe integration phase in a half sensing cycle such as 521 or 522. Insome embodiments, for example, the pre-charge/reset phase may be anorder of magnitude or more shorter than the integration phase in a halfsensing cycle such as 521 or 522. Further, as can be seen from FIG. 5,the reset phase is shorter than the pre-charge phase of FIG. 3,providing a higher sensing frequency. It should also be noted that insome embodiments, the pre-charge/reset phase illustrated in FIG. 5 issubstantially equal to or shorter than the reset phase oftranscapacitive sensing using the same sensor electrode. As the resetphase during transcapacitive sensing may be ⅕, 1/10, or less of thetranscapacitive integration phase, this means that the pre-charge/resetphase illustrated in FIG. 5 is substantially shorter than theconventional absolute sensing pre-charge phase illustrated in 320 ofFIG. 3. Further, in some embodiments, the combination ofpre-charge/reset phase and integration phase in a half sensing cycle(e.g., half sensing cycle 521 or 522) is substantially equal to orshorter than a half sensing cycle of transcapacitive sensing (e.g., 311or 312) accomplished using the same sensor electrode. Additionally, when2ΔV_(ref)>V_(dd)/2, the absolute capacitive sensing transmitter signalwhich drives the sensor electrode can have a higher modulation amplitudethan in the conventional technique for absolute capacitive sensingdescribed in relation to FIG. 3.

In one embodiment, as illustrated in FIG. 4B, the output is given byEquation 1.

$\begin{matrix}{V_{out} = {V_{ref} \mp \left( {{2\frac{C_{B}}{C_{FB}}\Delta\; V_{ref}} - {\Delta\; V_{ref}}} \right)}} & {{Equation}\mspace{14mu} 1}\end{matrix}$

As can be seen in Equation 1, the dynamic range of the output voltage islimited by the ΔV_(ref) term. In various embodiments, as can be seen inEquation 1, since last term is a signal which does not depend on C_(B),the capacitance to be measured, when C_(B)=0, V_(out≠)V_(ref). However,in Equation 1, when C_(B)=0 then V_(out)=V_(ref)±ΔV_(ref). Thus, in suchan embodiment, the system may be affine, but not linear, in C_(B) aroundV_(ref).

FIGS. 6A-6D illustrate operation of an capacitive measurement circuit600 through a full cycle of absolute capacitive sensing, according to anembodiment. Capacitance measurement circuit 600 may be included as partof an input device 100 and/or a processing system 110. For example,processing system 110 may supply input voltages for circuit 600 as wellas control signals which operate switches in circuit 600 and/or selectcapacitors from one or more banks of selectable capacitors. In FIGS.6A-6D the feedback capacitance C_(FB0) of FIGS. 4A and 4B has been splitinto multiple portions, C_(FB0) and C_(FB1), so that one portion C_(FB0)can be reset while the other portion C_(FB1) is pre-charged to either2V_(ref) or ground during the pre-charge/reset phase of a half sensingcycle. The selection of whether CFB1 is pre-charged to 2V_(ref) orground is accomplished by the positioning of switch SW3 during thepre-charge/reset phase of a half sensing cycle. In FIG. 6A-6D, theratios of feedback capacitors C_(FB0) and C_(FB1) may be chosen toprovide a charge integrator output, V_(out), which is a linear functionof the background capacitance, C_(B), while maintaining the same sensingfrequency increases as the method and circuit 400 shown in FIGS. 4A and4B. By using the pre-charged portion of C_(FB) as a charge subtractor,the embodiment of FIGS. 6A-6D can also handle larger values of C_(B)without saturating the receiver, increasing the dynamic range of thereceiver. For a given value of feedback capacitor C_(FB) from FIGS. 4Aand 4B, the capacitor is separated into two parallel capacitors C_(FB0)and C_(FB1). It should be appreciated that each of capacitors C_(FB0)and C_(FB1) may be composed of one or more selectable capacitorsselected from banks of selectable capacitances. As the total feedbackcapacitance is the sum of the capacitances of C_(FB0) and C_(FB1),feedback capacitor C_(FB) may be described as shown in Equation 2.C _(FB) =C _(FB0) +C _(FB1)  Equation 2

The ratio of these two capacitors may be chosen as a function of themodulation amplitude, ΔV_(ref), in order to substantially reduce theoffset term of Equation 1. In one embodiment, for each sensing cycle,there are two half cycles, with each half cycle having a reset phasefollowed by an integration phase. The integration phases are timeperiods where measurement takes place and may also be referred to asmeasurement phases.

During the first half cycle's reset phase, which is illustrated in FIG.6A as reset phase 1: switch SW1 is opened to decouple circuit 600 fromsensor electrode 270-0; C_(FB0) is discharged by closing switch SW2; themodulation of V_(AbsInput) on the non-inverting input of differentialamplifier 401 is shifted to V_(ref)+ΔV_(ref); and C_(FB1) is pre-chargedwith −(V_(ref)+ΔV_(ref))C_(FB1) coulombs by coupling switch SW3 with avoltage of 2V_(ref) (which may be at or near V_(dd) in someembodiments). The shift in reference voltage, V_(AbsInput), can occur atthe very beginning of the reset phase or at some time after the resetphase has started. Switch SW2 operates as a reset mechanism to dischargeand reset capacitor C_(FB0) additionally, switch SW3 also operates as areset mechanism by allowing capacitor C_(FB1) to be pre-charged andreset to a selected value. In some embodiments, when switch SW1 is open,sensor electrode 270-1 is left to electrically float and thussubstantially maintains whatever charge remained upon it.

During the first half cycle's integrate phase, which is illustrated inFIG. 6B as integrate phase 1: C_(FB0) and C_(FB1) are placed in parallelby coupling switch SW3 with the output of differential amplifier 401 andopening switch SW2; and the inverting input of differential amplifier401 is connected to C_(B) by closing switch SW1. This causes thepre-charge stored on C_(FB1) and any remaining charge to flow throughC_(FB0)+C_(FB1) to charge C_(B) with C_(B)(V_(ref)+ΔV_(ref)) coulombs.

Similar operations to those illustrated in FIGS. 6A and 6B take placeduring the second half-cycle, illustrated in FIGS. 6C and 6D, except thereference voltage of the amplifier, V_(AbsInput), is shifted toV_(ref)−ΔV_(ref) and C_(FB1) is pre-charged with(V_(ref)+ΔV_(ref))C_(FB1) coulombs.

During the second half cycle's reset phase, which is illustrated in FIG.6C as reset phase 2: switch SW1 is opened to decouple circuit 600 fromsensor electrode 270-0; C_(FB0) is discharged by closing switch SW2; themodulation of V_(AbsInput) on the non-inverting input of differentialamplifier 401 is shifted to V_(ref)−ΔV_(ref); and C_(FB1) is pre-chargedwith (V_(ref)+ΔV_(ref))C_(FB1) coulombs by coupling switch SW3 withground.

During the second half cycle's integrate phase, which is illustrated inFIG. 6D as integrate phase 2: C_(FB0) and C_(FB1) are placed in parallelby coupling switch SW3 with the output of differential amplifier 401 andopening switch SW2; and the inverting input of differential amplifier401 is connected to C_(B) by closing switch SW1. This causes thepre-charge stored on C_(FB1) and any remaining charge to flow throughC_(FB0)+C_(FB1) to charge C_(B) with C_(B)(V_(ref)−ΔV_(ref)) coulombs.

In one embodiment, C_(FB1) may be configured to increase the dynamicrange of the receiver channel by acting as a charge subtractor whichsubtracts charge from CB, and thus presents a smaller signal from sensorelectrode 270-0 for application by differential amplifier 401. Similarlystated, in various embodiments, C_(FB1) may be configured to increasethe range of C_(B) that may be measured by the receiver channel ofprocessing system 110 that is operating in absolute capacitance sensingmode. Further, C_(FB1) may be configured to increase the range of themodulation of the reference voltage for the receiver channel. Forexample, C_(FB1) may be configured to increase ΔV_(ref). Further, invarious embodiments, the value of C_(FB0) may be selected to achievedifferent levels of interference rejection. Further, in variousembodiments, the value of C_(FB0) may be selected to achieve differentlevels of gain (affecting signal to noise ratio (SNR), and sensitivity).In another embodiment, the value of C_(FB1) may be selected to achievedifferent levels of charge subtraction. In yet another embodiment, thevalues of C_(FB0) and C_(FB1) may be selected in concert with oneanother to achieve different levels of charge subtraction.

For the example shown in FIGS. 6A-6D, the V_(out) from differentialamplifier 401 may be given by Equation 3.

$\begin{matrix}{V_{out} = {V_{ref} \mp \left( {{2\frac{C_{B}}{C_{FB}}\Delta\; V_{ref}} + {\frac{C_{FB} - C_{{FB}\; 1}}{C_{FB}}\Delta\; V_{ref}} - {\frac{C_{{FB}\; 1}}{C_{FB}}V_{ref}}} \right)}} & {{Equation}\mspace{14mu} 3}\end{matrix}$

The modulation amplitude, a, may then be defined as shown in Equation 4.ΔV _(ref) =αV _(ref) where for some, 0<α≦1  Equation 4

Selection of α as shown in Equation 5 causes the last two terms ofEquation 3 to cancel out and the output is therefore represented moresimply, as shown in Equation 6.

$\begin{matrix}{\alpha = {\frac{C_{{FB}\; 1}}{C_{FB} - C_{{FB}\; 1}} = \frac{C_{{FB}\; 1}}{C_{{FB}\; 0}}}} & {{Equation}\mspace{14mu} 5} \\{V_{out} = {V_{ref} \mp {2\frac{C_{B}}{C_{FB}}\Delta\; V_{ref}}}} & {{Equation}\mspace{14mu} 6}\end{matrix}$

The output, V_(out), may be centered around V_(ref) and linear in C_(B).In various embodiments, for a known C_(FB) and α, Equation 5 can be usedto determine the values for C_(FB0) and C_(FB1). It should beappreciated that the simplification illustrated in Equation 5 is shownby way of example and not of limitation. That is to say, in otherembodiments, the simplified case of Equation 5 does not have to hold,and Equation 3 is not simplified in the manner demonstrated by Equation5.

FIG. 7 illustrates a capacitance measurement circuit 700, according toan embodiment. FIG. 7 illustrates an alternative implementation forabsolute capacitance sensing to the embodiments illustrated in FIGS.6A-6D and 4A and 4B. The implementation of FIG. 7 is similar to themethods described in relation to and depicted in FIGS. 6A-6D if C_(FB0)in those embodiments is set to a zero capacitance value or not includedand C_(FB1) is set to some non-zero value. The operation of circuit 700is the same as the operation of circuit 600, except that switch SW2 andcapacitor C_(FB0), and thus the entire C_(FB) (composed only of C_(FB1))is pre-charged during both reset phase 1 and reset phase 2. SW3 operatesas a reset mechanism by allowing capacitor C_(FB) to be pre-charged andreset to a selected value.

With circuit 700, as is similarly described above for circuits 400 and600, for each sensing cycle there are two half cycles, where each halfcycle has a reset phase followed by an integration phase. Theintegration phases are time periods where measurement takes place andmay also be referred to as measurement phases. In the first half cycle'sreset phase: switch SW1 is opened to decouple circuit 400 from sensorelectrode 270-1; C_(FB) is pre-charged by coupling switch SW3 with avoltage 2Vref; and the reference voltage, V_(AbsInput), of differentialamplifier 401 is shifted to V_(ref)+ΔV_(ref). The shift in referencevoltage, V_(AbsInput), can occur at the very beginning of the resetphase or at some time after the reset phase has started. In someembodiments, when switch SW1 is open, sensor electrode 270-1 is left toelectrically float and thus substantially maintains whatever chargeremained upon it. Then during the integrate phase of the first halfcycle: switch SW1 is closed to couple circuit 400 with sensor electrode270-0; switch SW3 is coupled with the output of differential amplifier401. These actions result in both C_(FB) and C_(B) being connected tothe differential amplifier 401 during this integrate phase. Thepre-charge stored on C_(FB) charges C_(B) with C_(B)(V_(ref)+ΔV_(ref))coulombs. A similar operation takes place in the second half-cycleexcept the operating point of the amplifier is shifted toV_(ref)−ΔV_(ref) and C_(FB) is pre-charged with (V_(ref)+ΔV_(ref))C_(FB)coulombs by positioning switch SW3 such that it is coupled with ground.

The integrator output, V_(out), for circuit 700 is given by Equation 7at the end of the first cycle integrate and second cycle integratephases, respectively.

$\begin{matrix}{V_{out} = \begin{Bmatrix}{2\frac{C_{B}}{C_{FB}}\Delta\; V_{ref}} \\{{2V_{ref}} - {2\frac{C_{B}}{C_{FB}}\Delta\; V_{ref}}}\end{Bmatrix}} & {{Equation}\mspace{14mu} 7}\end{matrix}$

In the embodiment illustrated in FIG. 7, the integrator output is alinear function of C_(B) while maintaining the same advantages asdescribed above in relation to FIGS. 4A-6D. In some embodiments ofcircuit 700, one or more techniques can also be employed to preventinadvertent saturation of differential amplifier 401 by peak transients.For example, in some embodiments one or more resistors can be added toattenuate peak transients on the inverting input to amplifier 401 toprevent charge loss through open switches and inadvertent saturation ofamplifier 401; in some embodiments, the timing of the operation ofswitches SW1 and SW3 can be adjusted such that for example switch SW1remains open during a transition from a reset phase to an integratephase until after switch SW3 is moved from either 2Vref or Ground tobeing coupled with V_(out) or else is switch SW1 is closed and thenswitch SW3 is repositioned from either 2V_(ref) or Ground to beingcoupled with V_(out). In some embodiments a combination of one or moreadded resistors and alterations in the timing of the operation ofswitches SW1 and SW3 may be utilized.

FIG. 8 illustrates a capacitance measurement circuit 400, according tosome embodiments. For example, circuit 400 of FIGS. 4A and 4B isillustrated in as being used as a charge integrator for transcapacitivesensing. This is an embodiment of the hybrid capacitive sensing devicethat was previously mentioned herein. In FIG. 8, a transmitter output,TX0 of processing system 110 drives a first sensor electrode (e.g.,transmitter electrode 260-0) with a transmitter signal, TX_(SIG). Atranscapacitance C_(TRANS) between transmitter electrode 260-0 andanother sensor electrode (e.g., receiver electrode 270-0) couples aresulting signal from this transmitter signal into this other sensorelectrode (e.g., receiver electrode 270-0). By supplying a fixed V_(ref)voltage on the non-inverting input of differential amplifier 401 andclosing switch SW1, this resulting signal can be received at input RX0(the inverting input of differential amplifier 401) and integrated bydifferential amplifier 401 so that the transcapacitive coupling ismeasured. It should be appreciated that circuits 600 and 700 may beoperated in a similar manner to perform transcapacitive sensing as partof a hybrid capacitive sensing circuit.

While many of the above embodiments are described in relation to adifferential amplifier 401 that configured as an integrating amplifier,similar approaches may be applied to embodiments comprising other typesof charge integrating techniques such as those comprising currentconveyors. Further, while in many embodiments a reset switch is employedto reset all or a portion of the feedback capacitance in the capacitivesensing circuit, in other embodiments, a resistor may be used. It shouldalso be appreciated that while the illustrated examples illustrate onlya single capacitive sensing circuit a processing system 110 may includenumerous capacitive sensing circuits, such as one per receiver channel.

Example Methods of Measuring Capacitance

FIGS. 9A-9C illustrate a flow diagram 900 of a method of capacitancemeasurement with a differential amplifier having an output anddifferential first and second input, according to various embodiments.In discussion of flow diagram 900 reference will be made to componentsand operates illustrated in one or more of FIGS. 6A-6D, FIGS. 4A-4B, andFIG. 7.

At procedure 910 of flow diagram 900, in one embodiment, a switchdisposed between a sensor electrode and a second input of a differentialamplifier is opened to initiate a reset phase where the sensor electrodeand the differential amplifier are decoupled. This is illustrated anddescribed in conjunction with circuits 400, 600, and 700 where switchSW1 is opened to decouple the inverting input of differential amplifier401 from sensor electrode 270-0.

At procedure 920 of flow diagram 900, in one embodiment, a feedbackcapacitance is reset to a first level of charge, the feedbackcapacitance is disposed between the second input and the output of thedifferential amplifier. The resetting can comprise a reset of all or aportion of the feedback capacitance. This resetting is illustrated anddescribed in conjunction with circuits 400 and 600 where switch SW2 incircuits 400 and 600 is closed to cause capacitive discharge and reset acapacitor (C_(FB) in circuit 400 or C_(FB0) in circuit 600) to a fullydischarged state. This resetting is also illustrated and described inconjunction with circuits 600 and 700 and reset phase 1, where switchSW3 in circuits 600 and 700 is positioned to cause pre-charge of acapacitor (C_(FB) in circuit 700 or C_(FB1) in circuit 600) to aselected value.

At procedure 930 of flow diagram 900, in one embodiment, the switch isclosed to initiate a measurement phase where the second input and thesensor electrode are coupled. This is illustrated and described inconjunction with circuits 400, 600, and 700 where switch SW1 is closedto couple the inverting input of differential amplifier 401 with sensorelectrode 270-0 and to facilitate an integration phase where thebackground capacitance associated with sensor electrode 270-0 isintegrated and thus measured.

Procedure 930 includes procedures 932 and 934. At procedure 932 of flowdiagram 900, in one embodiment, the measurement phase comprisesbalancing charge between the sensor electrode and the feedbackcapacitance such that a sensor electrode voltage equals a voltage of thefirst input equals a voltage of the second input, and the sensorelectrode is charged to a value proportional to its capacitance and thevoltage of the second input.

For example, with reference to circuits 400, 600, and 700, during thereset phase 1 the voltages on the non-inverting input, inverting input,and output of differential amplifier 401 balance and settle to the valueof V_(AbsInput) supplied to the non-inverting input. Closing switch SW1to initiate integration phase 1 (i.e., a measurement phase) causes thisbalancing to further act to charge sensor electrode 270-0. In themeasurement phase, sensor electrode 270-0 is connected to the secondinput (the inverting input) through switch SW1, and thus charge istransferred from/to the feedback capacitance (of circuits 400, 600, or700) to/from sensor electrode 270-0. The second input (the invertinginput) also settles to the same value as the voltage driven on the firstinput (the non-inverting input) of differential amplifier 401. The firstinput (the non-inverting input of differential amplifier 401) is drivenwith a modulated reference voltage. In this manner, the sensor electrodevoltage is equal to the voltage of second input of amplifier 401 whichis equal to the voltage applied to the first input of amplifier 401; andsensor electrode 270-0 is thus charged to a value proportional to itscapacitance and the voltage of the first and second inputs.

At procedure 934 of flow diagram 900, in one embodiment, the measurementphase also comprises utilizing the differential amplifier to integratecharge on the sensor electrode, such that an absolute capacitancecorresponding to a coupling between the sensor electrode and an inputobject is measured. For example, with reference to circuits 400, 600,and 700, by closing switch SW1, differential amplifier 401 integratesthe charge, C_(B), present on sensor electrode 270-0. C_(B) includes thebackground capacitance between the sensor electrode and ground, and ifan input object 140 is present in the sensing region associated withsensor electrode 270-0 its capacitance will be part of this backgroundcapacitance which is integrated and measured.

In some embodiments, the method as illustrated by procedures 910-930further comprises procedures 940 and 950.

At procedure 940 of flow diagram 900, in one embodiment, opening theswitch after the measurement phase to initiate a second occurrence ofthe reset phase. For example, as described in conjunction with circuits400, 600, and 700 sensing is split into two half cycles and each halfcycle includes a reset phase. Thus, if 910 describes the reset phase ofthe first half sensing cycle (e.g., 521 of FIG. 5), then 940 describesopening switch SW1 again to initiate the reset phase of the second halfsensing cycle (e.g., 522 of FIG. 5). During this reset phase the levelof modulation of V_(AbsInput) may be switched, such as fromV_(ref)+ΔV_(ref) to V_(ref)−ΔV_(ref).

At procedure 950, in one embodiment, the second reset phase involvesresetting the feedback capacitance to a second level of charge, whereinthe first and second levels of charge are different. With reference toFIG. 6B and FIG. 7, in one embodiment, this involves pre-charging all ora portion of a feedback capacitance to a different level of charge thanduring the first reset phase. This is accomplished by positioning ofswitch SW3. For, example if switch SW3 was coupled with 2V_(ref) duringthe first reset phase it would be coupled with ground during the secondreset phase.

In some embodiments, the method as illustrated by procedures 910-930further comprises procedure 960. At procedure 960 in one embodiment,during a transcapacitive sensing cycle, the differential amplifier isutilized to measure a resulting charge on the sensor electrode. Theresulting charge corresponds to a capacitive coupling between the sensorelectrode and a second sensor electrode, wherein the second sensorelectrode has been driven with a transmitter signal. This is illustratedin FIG. 8 which shows differential amplifier 401 being used to integratea resulting charge/signal on receiver electrode 270-0, where thatresulting charge results from transmitter electrode 260-0 being drivenwith a transmitter signal by processing system 110.

The examples set forth herein were presented in order to best explain,to describe particular applications, and to thereby enable those skilledin the art to make and use embodiments of the described examples.However, those skilled in the art will recognize that the foregoingdescription and examples have been presented for the purposes ofillustration and example only. The description as set forth is notintended to be exhaustive or to limit the embodiments to the preciseform disclosed.

What is claimed is:
 1. A method of capacitance measurement with adifferential amplifier having an output and differential first andsecond inputs, said method comprising: opening a switch disposed betweena sensor electrode and said second input of said differential amplifierto initiate a reset phase where said sensor electrode and saiddifferential amplifier are decoupled; resetting a feedback capacitanceto a first level of charge, said feedback capacitance disposed betweensaid second input and said output; closing said switch to initiate ameasurement phase where said second input and said sensor electrode arecoupled, said measurement phase comprising; balancing charge betweensaid sensor electrode and said feedback capacitance such that a sensorelectrode voltage equals a voltage of said first input equals a voltageof said second input, and said sensor electrode is charged to a valueproportional to its capacitance and said voltage of said second input;and utilizing said differential amplifier to integrate charge on saidsensor electrode, such that an absolute capacitance is measured; openingsaid switch after said measurement phase to initiate a second occurrenceof said reset phase; and resetting said feedback capacitance to a secondlevel of charge during said second occurrence of said reset phase,wherein said first and second levels of charge are different.
 2. Themethod as recited in claim 1, further comprising: during atranscapacitive sensing cycle, utilizing said differential amplifier tomeasure a resulting charge on said sensor electrode, said resultingcharge corresponding to a capacitive coupling between said sensorelectrode and a second sensor electrode, wherein said second sensorelectrode has been driven with a transmitter signal.
 3. A capacitancemeasurement circuit comprising: a differential amplifier comprising:differential first and second inputs; and an output; a switch coupledwith said second input, said switch having a closed state and an openstate, wherein said second input is coupled with a sensor electrode in ameasurement phase when said switch is in said closed state, and whereinsaid second input is decoupled with said sensor electrode in a resetphase when said switch is in said open state; a feedback capacitancecoupled between said output and said second input; a reset mechanismcoupled in parallel with at least a portion of said feedback capacitanceand configured to reset said feedback capacitance to a first level ofcharge during a first occurrence of said reset phase, wherein said resetmechanism is further configured to reset said feedback capacitance to asecond level of charge during a second occurrence of said reset phase,wherein said first and second levels of charge are different; andwherein during said measurement phase said differential amplifieroperates to charge said sensor electrode while balancing voltages onsaid first and second inputs to a voltage level associated with amodulated reference voltage coupled with said first input and tointegrate charge on said sensor electrode to measure capacitancecorresponding to a coupling between said sensor electrode and an inputobject.
 4. The circuit of claim 3, wherein said feedback capacitance isconfigured to be pre-charged during said reset phase.
 5. The circuit ofclaim 3, wherein at least a portion of said feedback capacitance acts asa charge subtractor which increases a dynamic range of said differentialamplifier.
 6. The circuit of claim 3, wherein said reset phase isshorter than said measurement phase.
 7. The circuit of claim 6, whereinsaid reset phase is at least an order of magnitude shorter than saidmeasurement phase.
 8. The circuit of claim 3, wherein said voltage levelof said modulated reference voltage varies in different operating modesof said circuit.
 9. The circuit of claim 3, wherein said measurementphase is an absolute capacitance measurement phase.
 10. The circuit ofclaim 3, wherein said feedback capacitance is formed by a plurality ofselectable capacitors.
 11. An input device comprising: a first sensorelectrode; a second sensor electrode; a transmitter coupled with saidsecond sensor electrode and configured to drive a transmitter signal onsaid second sensor electrode; a differential amplifier comprising:differential first and second inputs; and an output; a switch coupledwith said second input, said switch having a closed state and an openstate, wherein said second input is coupled with said first sensorelectrode in a measurement phase when said switch is in said closedstate, and wherein said second input is decoupled with said first sensorelectrode in a reset phase when said switch is in said open state; afeedback capacitance coupled between said output and said second input;a reset mechanism coupled in parallel with at least a portion of saidfeedback capacitance and configured to reset said feedback capacitanceto a first level of charge during a first occurrence of said resetphase; and wherein, during said measurement phase, said differentialamplifier operates to charge said first sensor electrode while balancingvoltages on said first and second inputs to a voltage level associatedwith a modulated reference voltage coupled with said first input and tointegrate charge on said first sensor electrode to measure capacitancecorresponding to a coupling between said sensor electrode and an inputobject; and wherein, during a transcapacitive sensing cycle of saidinput device, said differential amplifier is further configured tomeasure a resulting charge on said first sensor electrode correspondingto a capacitive coupling between said first and second sensorelectrodes.
 12. The input device of claim 11, wherein time lengths ofsaid reset phase and a transcapacitive reset phase of saidtranscapacitive sensing cycle are substantially equal.
 13. The inputdevice of claim 11, wherein said reset phase is shorter than saidmeasurement phase.
 14. The input device of claim 11, wherein saidmodulated reference voltage comprises a waveform which is symmetricabout a reference voltage.
 15. The input device of claim 11, whereinsaid reset mechanism is further configured to reset said feedbackcapacitance to a second level of charge during a second occurrence ofsaid reset phase, wherein said first and second levels of charge aredifferent.
 16. The input device of claim 11, wherein said feedbackcapacitance is configured to be pre-charged during said reset phase. 17.The input device of claim 11, wherein at least a portion of saidfeedback capacitance acts as a charge subtractor which increases adynamic range of said differential amplifier.
 18. The input device ofclaim 11, wherein said voltage level of said modulated reference voltagevaries in different operating modes of said input device.
 19. The inputdevice of claim 11, wherein said measurement phase is an absolutecapacitance measurement phase.
 20. The input device of claim 11, whereinsaid feedback capacitance is formed by a plurality of selectablecapacitors.
 21. The input device of claim 11, wherein said first sensorelectrode electrically floats during said reset phase.
 22. The inputdevice of claim 11, wherein said modulated reference voltage transitionsat a beginning of said reset phase.
 23. A method of capacitancemeasurement with a differential amplifier having an output anddifferential first and second inputs, said method comprising: during anabsolute capacitive sensing cycle: opening a switch disposed between asensor electrode and said second input of said differential amplifier toinitiate a reset phase where said sensor electrode and said differentialamplifier are decoupled; resetting a feedback capacitance to a firstlevel of charge, said feedback capacitance disposed between said secondinput and said output; closing said switch to initiate a measurementphase where said second input and said sensor electrode are coupled,said measurement phase comprising; balancing charge between said sensorelectrode and said feedback capacitance such that a sensor electrodevoltage equals a voltage of said first input equals a voltage of saidsecond input, and said sensor electrode is charged to a valueproportional to its capacitance and said voltage of said second input;and utilizing said differential amplifier to integrate charge on saidsensor electrode, such that an absolute capacitance is measured; andduring a transcapacitive sensing cycle: utilizing said differentialamplifier to measure a resulting charge on said sensor electrode, saidresulting charge corresponding to a capacitive coupling between saidsensor electrode and a second sensor electrode, wherein said secondsensor electrode has been driven with a transmitter signal.
 24. Themethod as recited in claim 23, further comprising: during said absolutecapacitive sensing cycle: opening said switch after said measurementphase to initiate a second occurrence of said reset phase; and resettingsaid feedback capacitance to a second level of charge during said secondoccurrence of said reset phase, wherein said first and second levels ofcharge are different.